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1 LT1512 sepic constant-current/ constant-voltage battery charger the lt ? 1512 is a 500khz current mode switching regula- tor specially configured to create a constant-current/ constant-voltage battery charger. in addition to the usual voltage feedback node, it has a current sense feedback circuit for accurately controlling output current of a flyback or sepic (single-ended primary inductance converter) topology charger. these topologies allow the current sense circuit to be ground referred and completely sepa- rated from the battery itself, simplifying battery switching and system grounding problems. in addition, these to- pologies allow charging even when the input voltage is lower than the battery voltage. maximum switch current on the LT1512 is 1.5a. this allows battery charging currents up to 1a for a single lithium-ion cell. accuracy of 1% in constant-voltage mode is perfect for lithium battery applications. charging cur- rent can be easily programmed for all battery types. descriptio n u applicatio n s u n battery charging of nicd, nimh, lead-acid or lithium rechargeable cells n precision current limited power supply n constant-voltage/constant-current supply n transducer excitation *maximum input voltage = 40v C v bat n charger input voltage may be higher, equal to or lower than battery voltage n charges any number of cells up to 30v* n 1% voltage accuracy for rechargeable lithium batteries n 100mv current sense voltage for high efficiency n battery can be directly grounded n 500khz switching frequency minimizes inductor size n charging current easily programmable or shut down features , ltc and lt are registered trademarks of linear technology corporation. typical applicatio n u figure 1. sepic charger with 0.5a output current maximum charging current actual programmed charging current will be independent of input voltage and battery voltage if it does not exceed the values shown. these are electrical limitations based on maximum switch current. package thermal limitations may reduce maximum charging current. see applications information. input voltage (v) 0 current (a) 0.6 0.8 1.0 20 1512 ta02 0.4 0.2 0 5 10 15 25 inductor = 33 m h double lithium cell (8.2v) 6v battery 12v battery single lithium cell (4.1v) LT1512 i fb v c v in l1 a* l1 b* 0.5a gnd gnd s fb 1512 f01 v sw sync and/or shutdown wall adapter input s/s c3 22 m f 25v c2** 2.2 m f c5 0.1 m f * ** l1 a, l1 b are two 33 m h windings on a single inductor: coiltronics ctx33-3 tokin ceramic 1e225zy5u-c203-f c4 0.22 m f r4 24 w + charge shutdown r1 r2 r3 0.2 w r5 1k c1 22 m f 25v + d1 mbrs130lt3
2 LT1512 a u g w a w u w a r b s o lu t exi t i s wu u package / o rder i for atio symbol parameter conditions min typ max units v ref v fb reference voltage measured at fb pin 1.233 1.245 1.257 v v c = 0.8v l 1.228 1.245 1.262 v fb input current v fb = v ref 300 550 na l 600 na fb reference voltage line regulation 2.7v v in 25v, v c = 0.8v l 0.01 0.03 %/v v iref i fb reference voltage measured at i fb pin C 107 C 100 C 93 mv v fb = 0v, v c = 0.8v l C110 C 100 C 90 mv i fb input current v ifb = v iref (note 2) l 10 25 35 m a i fb reference voltage line regulation 2.7v v in 25v, v c = 0.8v l 0.01 0.05 %/v g m error amplifier transconductance d i c = 25 m a 1100 1500 1900 m mho l 700 2300 m mho error amplifier source current v fb = v ref C 150mv, v c = 1.5v l 120 200 350 m a error amplifier sink current v fb = v ref + 150mv, v c = 1.5v l 1400 2400 m a error amplifier clamp voltage high clamp, v fb = 1v 1.70 1.95 2.30 v low clamp, v fb = 1.5v 0.25 0.40 0.52 v a v error amplifier voltage gain 500 v/ v v c pin threshold duty cycle = 0% 0.8 1 1.25 v f switching frequency 2.7v v in 25v 450 500 550 khz 0 c t j 125 c l 430 500 580 khz C40 c t j < 0 c (LT1512i) 400 580 khz maximum switch duty cycle l 88 95 % switch current limit blanking time 130 260 ns bv output switch breakdown voltage 0 c t j 125 c l 40 47 v C40 c t j < 20 c (LT1512i) 35 v v in = 5v, v c = 0.6v, v fb = v ref , i fb = 0v, v sw and s/s pins open, unless otherwise noted. e lectr ic al c c hara terist ics input voltage .......................................................... 30v switch voltage ........................................................ 40v s/s pin voltage ....................................................... 30v fb pin voltage (transient, 10ms) ......................... 10v v fb pin current .................................................... 10ma i fb pin voltage (transient, 10ms) ......................... 10v storage temperature range ................ C 65 c to 150 c ambient temperature range LT1512c (note 3) .................................... 0 c to 70 c LT1512i .............................................. C 40 c to 85 c operating junction temperature range LT1512c (note 3) ............................ C 20 c to 125 c LT1512i ............................................ C 40 c to 125 c short circuit ......................................... 0 c to 150 c lead temperature (soldering, 10 sec)................. 300 c order part number t jmax = 125 c, q ja = 100 c/ w (n) t jmax = 125 c, q ja = 130 c/ w (s) consult factory for military grade parts. 1 2 3 4 8 7 6 5 top view v c fb i fb s/s v sw gnd gnd s v in n8 package 8-lead pdip s8 package 8-lead plastic so s8 part marking 1512 1512i LT1512cn8 LT1512cs8 LT1512in8 LT1512is8 note: contact factory concerning 16-lead fused-lead gn package with lower thermal resistance
3 LT1512 symbol parameter conditions min typ max units v sat output switch on resistance i sw = 2a l 0.5 0.8 w i lim switch current limit duty cycle = 50% l 1.5 1.9 2.7 a duty cycle = 80% (note 1) l 1.3 1.7 2.5 a d i in supply current increase during switch on time 15 25 ma/a d i sw control voltage to switch current 2a/v transconductance minimum input voltage l 2.4 2.7 v i q supply current 2.7v v in 25v l 4 5.5 ma shutdown supply current 2.7v v in 25v, v s/s 0.6v 0 c t j 125 c l 12 30 m a C40 c t j 0 c (LT1512i) 50 m a shutdown threshold 2.7v v in 25v l 0.6 1.3 2 v shutdown delay l 51225 m s s/s pin input current 0v v s/s 5v l C10 15 m a synchronization frequency range l 600 800 khz e lectr ic al c c hara terist ics v in = 5v, v c = 0.6v, v fb = v ref , i fb = 0v, v sw and s/s pins open, unless otherwise noted. the l denotes specifications which apply over the full operating temperature range. note 1: for duty cycles (dc) between 50% and 85%, minimum guaranteed switch current is given by i lim = 0.667 (2.75 C dc). note 2: the i fb pin is servoed to its regulating state with v c = 0.8v. note 3: commercial devices are guaranteed over 0 c to 125 c junction temperature range and 0 c to 70 c ambient temperature range. these parts are also designed, characterized and expected to operate over the C20 c to 85 c extended ambient temperature range, but are not tested at C20 c or 85 c. devices with full guaranteed electrical specifications over the ambient temperature range C 40 c to 85 c are available as industrial parts with an i suffix. maximum allowable ambient temperature may be limited by power dissipation. parts may not necessarily be operated simultaneously at maximum power dissipation and maximum ambient temperature. temperature rise calculations must be done as shown in the applications information section to ensure that maximum junction temperature does not exceed 125 c limit. with high power dissipation, maximum ambient temperature may be less than 70 c. typical perfor m a n ce characteristics u w switch saturation voltage vs switch current minimum input voltage vs temperature switch current limit vs duty cycle switch current (a) 0 switch saturation voltage (v) 0.6 0.8 1.0 1.6 1512 g01 0.4 0.2 0.5 0.7 0.9 0.3 0.1 0 0.4 0.8 1.2 2.0 1.4 0.2 0.6 1.0 1.8 100 c 150 c 25 c ?5 c duty cycle (%) switch current limit (a) 3.0 2.5 2.0 1.5 1.0 0.5 0 20 40 60 80 1512 g02 100 10 0 30 50 70 90 25 c and 125 c ?5 c temperature ( c) ?0 1.8 input voltage (v) 2.0 2.2 2.4 2.6 050 100 150 1512 g03 2.8 3.0 ?5 25 75 125
4 LT1512 typical perfor m a n ce characteristics u w minimum peak-to-peak synchronization voltage vs temp feedback input current vs temperature negative feedback input current vs temperature temperature ( c) ?0 0 minimum synchronization voltage (v p-p ) 0.5 1.0 1.5 2.0 050 100 150 1512 g04 2.5 3.0 ?5 25 75 125 f sync = 700khz temperature ( c) ?0 feedback input current (na) 400 500 600 150 1512 g05 300 200 0 0 50 100 100 800 700 ?5 25 75 125 v fb = v ref temperature ( c) ?0 ?0 negative feedback input current ( m a) ?0 0 0 50 75 1512 g06 ?0 ?0 ?0 ?5 25 100 125 150 pi n fu n ctio n s uuu v c : the compensation pin is primarily used for frequency compensation, but it can also be used for soft starting and current limiting. it is the output of the error amplifier and the input of the current comparator. peak switch current increases from 0a to 1.8a as the v c voltage varies from 1v to 1.9v. current out of the v c pin is about 200 m a when the pin is externally clamped below the internal 1.9v clamp level. loop frequency compensation is performed with a capacitor or series rc network from the v c pin directly to the ground pin (avoid ground loops). fb: the feedback pin is used for positive output voltage sensing. this pin is the inverting input to the voltage error amplifier. the r1/r2 voltage divider connected to fb defines li-ion float voltage at full charge, or acts as a voltage limiter for nicd or nimh applications. input bias current is typically 300na, so divider current is normally set to 100 m a to swamp out any output voltage errors due to bias current. the noninverting input of this amplifier is tied internally to a 1.245v reference. the grounded end of the output voltage divider should be connected directly to the LT1512 ground pin (avoid ground loops). i fb : the current feedback pin is used to sense charging current. it is the input to a current sense amplifier that controls charging current when the battery voltage is below the programmed voltage. during constant-current operation, the i fb pin regulates at C 100mv. input resis- tance of this pin is 5k w , so filter resistance (r4, figure 1) should be less than 50 w . the 24 w , 0.22 m f filter shown in figure 1 is used to convert the pulsating current in the sense resistor to a smooth dc current feedback signal. s/s: this pin can be used for shutdown and/or synchroni- zation. it is logic level compatible, but can be tied to v in if desired. it defaults to a high on state when floated. a logic low state will shut down the charger to a micropower state. driving the s/s pin with a continuous logic signal of 600khz to 800khz will synchronize switching frequency to the external signal. shutdown is avoided in this mode with an internal timer. v in : the input supply pin should be bypassed with a low esr capacitor located right next to the ic chip. the grounded end of the capacitor must be connected directly to the ground plane to which the gnd pin is connected. gnd s, gnd: the LT1512 uses separate ground pins for switch current (gnd) and the control circuitry (gnd s). this isolates the control ground from any induced voltage created by fast switch currents. both pins should be tied directly to the ground plane, but the external control circuit components such as the voltage divider, frequency compensation network and i fb bypass capacitor should
5 LT1512 pi n fu n ctio n s uuu be connected directly to the gnd s pin or to the ground plane close to the point where the gnd s pin is connected. v sw : the switch pin is the collector of the power switch, carrying up to 1.5a of current with fast rise and fall times. keep the traces on this pin as short as possible to minimize radiation and voltage spikes. in particular, the path in figure 1 which includes sw to c2, d1, c1 and around to the LT1512 ground pin should be as short as possible to minimize voltage spikes at switch turn-off. + i fba i fb s/s fb 5k 62k 0.08 w + ea v c v in gnd 1512 f02 gnd s 1.245v ref 500khz osc sync shutdown delay and reset low dropout 2.3v reg anti-sat logic driver sw switch + ia a v ? 6 comp figure 2 block diagra m w operatio n u the LT1512 is a current mode switcher. this means that switch duty cycle is directly controlled by switch current rather than by output voltage or current. referring to the block diagram, the switch is turned on at the start of each oscillator cycle. it is turned off when switch current reaches a predetermined level. control of output voltage and current is obtained by using the output of a dual feedback voltage sensing error amplifier to set switch current trip level. this technique has the advantage of simplified loop frequency compensation. a low dropout internal regulator provides a 2.3v supply for all internal circuitry on the LT1512. this low dropout design allows input voltage to vary from 2.7v to 25v. a 500khz oscillator is the basic clock for all internal timing. it turns on the output switch via the logic and driver circuitry. special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. this minimizes driver dissipation and provides very rapid turn-off of the switch. a unique error amplifier design has two inverting inputs which allow for sensing both output voltage and current. a 1.245v bandgap reference biases the noninverting input. the first inverting input of the error amplifier is brought out for positive output voltage sensing. the second inverting input is driven by a current amplifier which is sensing output current via an external current sense resistor. the
6 LT1512 operatio n u current amplifier is set to a fixed gain of C 12.5 which provides a C 100mv current limit sense voltage. the error signal developed at the amplifier output is brought out externally and is used for frequency compen- sation. during normal regulator operation this pin sits at a voltage between 1v (low output current) and 1.9v (high output current). switch duty cycle goes to zero if the v c pin is pulled below the v c pin threshold, placing the LT1512 in an idle mode. the LT1512 is an ic battery charger chip specifically opti- mized to use the sepic converter topology. the sepic topology has unique advantages for battery charging. it will operate with input voltages above, equal to or below the battery voltage, has no path for battery discharge when turned off and eliminates the snubber losses of flyback designs. it also has a current sense point that is ground referred and need not be connected directly to the battery. the two inductors shown are actually just two identical windings on one inductor core, although two separate inductors can be used. a current sense voltage is generated with respect to ground across r3 in figure 1. the average current through r3 is always identical to the current delivered to the battery. the LT1512 current limit loop will servo the voltage across r3 to C 100mv when the battery voltage is below the voltage limit set by the output divider r1/r2. constant current charging is therefore set at 100mv/r3. r4 and c4 filter the current signal to deliver a smooth feedback voltage to the i fb pin. r1 and r2 form a divider for battery voltage sensing and set the battery float voltage. the suggested value for r2 is 12.4k. r1 is calculated from: r rv ra bat 1 2 1 245 1 245 2 0 3 = +m (.) .(.) v bat = battery float voltage 0.3 m a = typical fb pin bias current a value of 12.4k for r2 sets divider current at 100 m a. this is a constant drain on the battery when power to the charger is off. if this drain is too high, r2 can be increased to 41.2k, reducing divider current to 30 m a. this introduces an addi- applicatio n s i n for m atio n wu u u tional uncorrectable error to the constant voltage float mode of about 0.5% as calculated by: v error = 0.15 a(r1)(r2) 1.245(r1+ r2) bat m 0.15 m a = expected variation in fb bias current around the nominal 0.3 m a typical value. with r2 = 41.2k and r1 = 228k, (v bat = 8.2v), the error due to variations in bias current would be 0.42%. a second option is to disconnect the voltage divider with a small nmos transistor as shown in figure 3. to ensure adequate drive to the transistor (even when the v in voltage is at its lowest operating point of 2.4v), the fet gate is driven wth a peak detected voltage via d2. note that there are two connections for d2. the l1 a connection must be used if the voltage divider is set for less than 3.5v (fully charged battery). gate drive is equal to battery voltage plus input voltage. the disadvantage of this connection is that q1 will still be on if the v in voltage is active and the charger is shut down via the s/s pin. the l1 b connection allows q1 to turn off when v in is off or when shutdown is initiated, but the reduced gate drive (=v bat ) is not adequate to ensure a q1 on-state for fully charged battery voltages less than 3.5v. do not substitute for q1 unless the new device has adequate v gs maximum rating, especially if d2 is connected to l1a. c6 filters the gate drive and r5 pulls the gate low when switching stops. disconnecting the divider leaves only d1 diode leakage as a battery drain. see diode selection for a discussion of diode leakage.
7 LT1512 applicatio n s i n for m atio n wu u u LT1512 v in l1 a l1 b gnd fb 1512 f03 v sw shutdown d2 1n4148 connect d2 anode here if fully charged battery voltage is greater than 3.5v and q1 must be turned off in shutdown with v in still active connect d2 anode here for fully charged battery voltage less than 3.5v. q1 will not be turned off in shutdown if v in is present s/s c2 c6 470pf r1 r5 470k r2 r3 q1 2n7002 + d1 figure 3. eliminating divider current maximum input voltage maximum input voltage for the circuit in figure 1 is partly determined by battery voltage. a sepic converter has a maximum switch voltage equal to input voltage plus output voltage. the LT1512 has a maximum input voltage of 30v and a maximum switch voltage of 40v, so this limits maximum input voltage to 30v, or 40v C v bat , whichever is less. maximum v bat = 40v C v in . shutdown and synchronization the dual function s/s pin provides easy shutdown and synchronization. it is logic level compatible and can be pulled high or left floating for normal operation. a logic low on the s/s pin activates shutdown, reducing input supply current to 12 m a. to synchronize switching, drive the s/s pin between 600khz and 800khz. inductor selection l1a and l1b are normally just two identical windings on one core, although two separate inductors can be used. a typical value is 33 m h, which gives about 0.25a peak-to-peak induc- tor current. lower values will give higher ripple current, which reduces maximum charging current. 15 m h can be used if charging currents are at least 20% lower than the values shown in the maximum charging current graph. higher inductance values give slightly higher maximum charging current, but are larger and more expensive. a low loss toroid core such as koolm m ? , molypermalloy or metglas ? is recommended. series resistance should be less than 0.1 w for each winding. open core inductors, such as rods or barrels are not recommended because they generate large magnetic fields which may interfere with other elec- tronics close to the charger. input capacitor the sepic topology has relatively low input ripple current compared to other topologies and higher harmonics are especially low. rms ripple current in the input capacitor is less than 0.1a with l = 33 m h and less than 0.2a with l = 15 m h. a low esr 22 m f, 25v solid tantalum capacitor (avx type tps or sprague type 593d) is adequate for most applications with the following caveat. solid tantalum ca- pacitors can be destroyed with a very high turn-on surge current such as would be generated if a low impedance input source were hot switched to the charger input. if this condition can occur, the input capacitor should have the highest possible voltage rating, at least twice the surge input voltage if possible. consult with the capacitor manufacturer before a final choice is made. a 2.2 m f ceramic capacitor such as the one used for the coupling capacitor can also be used. these capacitors do not have a turn-on surge limitation. the input capacitor must be connected directly to the v in pin and the ground plane close to the LT1512. koolm m is a registered trademark of magnetics, inc. metglas is a registered trademark of alliedsignal inc.
8 LT1512 applicatio n s i n for m atio n wu u u output capacitor it is assumed as a worst case that all the switching output ripple current from the battery charger could flow in the output capacitor. this is a desirable situation if it is neces- sary to have very low switching ripple current in the battery itself. ferrite beads or line chokes are often inserted in series with the battery leads to eliminate high frequency currents that could create emi problems. this forces all the ripple current into the output capacitor. total rms current into the capacitor has a maximum value of about 0.5a, and this is handled with a 22 m f, 25v capacitor shown in figure 1. this is an avx type tps or sprague type 593d surface mount solid tantalum unit intended for switching applications. do not substitute other types without ensuring that they have adequate ripple current ratings. see input capacitor section for details of surge limitation on solid tantalum capacitors if the battery may be hot switched to the output of the charger. coupling capacitor c2 in figure 1 is the coupling capacitor that allows a sepic converter topology to work with input voltages either higher or lower than the battery voltage. dc bias on the capacitor is equal to input voltage. rms ripple current in the coupling capacitor has a maximum value of about 0.5a at full charg- ing current. a conservative formula to calculate this is: i ivv v coup rms chrg in bat in () ()(.) () = + 11 2 (1.1 is a fudge factor to account for inductor ripple current and other losses) with i chrg = 0.5a, v in = 15v and v bat = 8.2v, i coup = 0.43a the recommended capacitor is a 2.2 m f ceramic type from marcon or tokin. these capacitors have extremely low esr and high ripple current ratings in a small package. solid tantalum units can be substituted if their ripple current rating is adequate, but typical values will increase to 22 m f or more to meet the ripple current requirements. diode selection the switching diode should be a schottky type to minimize both forward and reverse recovery losses. average diode current is the same as output charging current , so this will be under 1a. a 1a diode is recommended for most applica- tions, although smaller devices could be used at reduced 1 2 3 4 v c fb i fb s/s v sw gnd gnd s v in r5 c4 r3 r4 s/s 1512 f04b c5 r2 c1 d1 c2 c3 r1 v batt + + +v in gnd l1b l1a r4 b. single-sided altenative layout 2 winding inductor l1a 1512 f04a l1b r4 r1 1 4 3 2 c4 r2 d1 v in gnd gnd v batt r3 c2a c2b c3 c5 c1 u1 r5 s/s a. double-sided (vias connect to the backside of ground plane. dash lines indicate interconnects on backside. demo board uses this layout, except that r5 has been added to increase phase margin) figure 4. LT1512 suggested layouts for critical thermal and electrical paths
9 LT1512 charging current. maximum diode reverse voltage will be equal to input voltage plus battery voltage. diode reverse leakage current will be of some concern during charger shutdown. this leakage current is a direct drain on the battery when the charger is not powered. high current schottky diodes have relatively high leakage cur- rents (2 m a to 200 m a) even at room temperature. the latest very-low-forward devices have especially high leakage cur- rents. it has been noted that surface mount versions of some schottky diodes have as much as ten times the leakage of their through-hole counterparts. this may be because a low forward voltage process is used to reduce power dissipation in the surface mount package. in any case, check leakage specifications carefully before making a final choice for the switching diode. be aware that diode manufacturers want to specify a maximum leakage current that is ten times higher than the typical leakage. it is very difficult to get them to specify a low leakage current in high volume production. this is an on going problem for all battery charger circuits and most customers have to settle for a diode whose typical leakage is adequate, but theoretically has a worst-case condition of higher than desired battery drain. thermal considerations care should be taken to ensure that worst-case conditions do not cause excessive die temperatures. typical thermal resistance is 130 c/ w for the s8 package but this number will vary depending on the mounting technique (copper area, air flow, etc). average supply current (including driver current) is: ima vi v in bat chrg in =+ 4 0 024 ()( )(.) switch power dissipation is given by: p irvvv v sw chrg sw bat in bat in = + ()()( )() () 2 2 r sw = output switch on resistance total power dissipation of the die is equal to supply current times supply voltage, plus switch power: p d(total) = (i in )(v in ) + p sw applicatio n s i n for m atio n wu u u for v in = 10v, v bat = 8.2v, i chrg = 0.5a, r sw = 0.65 w i in = 4ma + 10ma = 14ma p sw = 0.24w p d = (0.014)(10) + 0.24 = 0.38w the s8 package has a thermal resistance of 130 c/w. (contact factory concerning 16-lead fused-lead package with footprint approximately same as s8 package and with lower thermal resistance.) die temperature rise will be (0.38w)(130 c/w) = 49 c. a maximum ambient tempera- ture of 60 c will give a die temperature of 60 c + 49 c = 109 c. this is only slightly less than the maximum junction temperature of 125 c, illustrating the importance of doing these calculations! programmed charging current LT1512 charging current can be programmed with a pwm signal from a processor as shown in figure 5. c6 and d2 form a peak detector that converts a positive logic signal to a negative signal. the average negative signal at the input to r5 is equal to the processor v cc level multiplied by the inverse pwm ratio. this assumes that the pwm signal is a cmos output that swings rail-to-rail with a source resis- tance less than a few hundred ohms. the negative voltage is converted to a current by r5 and r6 and filtered by c7. this current multiplied by r4 generates a voltage that subtracts from the 100mv sense voltage of the LT1512. this is not a high precision technique because of the errors in v cc and the diode voltage, but it can typically be used to adjust charging current over a 20% to 100% range with good repeatability (full charging current accuracy is not affected). to reduce the load on the logic signal, r4 has been increased + c6 1 m f c7 10 m f c4 0.22 m f r3 1512 f05 l1b i fb LT1512 r5 4.02k pwm input 3 1khz d2 r6 4.02k r4 200 w + figure 5. programming charge current
10 LT1512 applicatio n s i n for m atio n wu u u from 24 w to 200 w . this causes a known increase in full- scale charging current (pwm = 0) of 3% due to the 5k input resistance of the i fb pin. note that 100% duty cycle gives full charging current and that very low duty cycles (especially zero!) will not operate correctly. very low duty cycle (<10%) is a problem because the peak detector requires a finite up-time to reset c6. more help linear technology field application engineers have a cad spreadsheet program for detailed calculations of circuit operating conditions, and our applications department is always ready to lend a helping hand. for additional informa- tion refer to the lt1372 data sheet. this part is identical to the LT1512 except for the current amplifier circuitry. dimensions in inches (millimeters) unless otherwise noted. package descriptio n u n8 package 8-lead pdip (narrow 0.300) (ltc dwg # 05-08-1510) n8 0695 0.005 (0.127) min 0.100 0.010 (2.540 0.254) 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 0.005 (3.302 0.127) 0.015 (0.380) min 0.018 0.003 (0.457 0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.255 0.015* (6.477 0.381) 0.400* (10.160) max 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.025 0.015 +0.635 0.381 8.255 () *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.010 inch (0.254mm)
11 LT1512 dimensions in inches (millimeters) unless otherwise noted. package descriptio n u information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 0695 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
12 LT1512 ? linear technology corporation 1996 lt/gp 1096 7k ? printed in the usa linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax : (408) 434-0507 l telex : 499-3977 typical applicatio n u the circuit in figure 6 will provide adapter current limiting to ensure that the battery charger never overloads the adapter. in addition, it adjusts charging current to a lower value if other system power increases to the point where the adapter would be overloaded. this allows the LT1512 to charge the battery at the maximum possible rate without concern about varying system power levels. the lm301 op amp used here is unusual in that it can operate with its inputs at a voltage equal to the positive supply voltage. LT1512 i fb v c v in l1 a* l1 b* 0.5a 1 1 3 3 2 2 8 8 5 4 4 6 7 7 6 to fb pin d2 1n4148 gnd gnd s fb 1512 f06 v sw sync and/or shutdown wall adapter input s/s c3 22 m f 25v system power c2** 2.2 m f c5 0.1 m f r5 1k r7 12k q1 2n3904 * ** l1 a, l1 b are two 33 m h windings on a common core: coiltronics ctx33-3 tokin ceramic 1e225zy5u-c203-f c4 0.22 m f r4 24 w + r1 r2 r3 0.2 w c1 22 m f 25v + d1 mbrs130lt3 r6 0.2 w + 30pf lm301 figure 6. adding adapter current limiting part number description comments lt1239 backup battery management system charges backup battery and regulates backup battery output when main battery removed ltc ? 1325 microprocessor controlled battery management system can charge, discharge and gas gauge nicd, nimh and pb-acid batteries with software charging profiles lt1510 1.5a constant-current/constant-voltage battery charger step-down charger for li-ion, nicd and nimh lt1511 3.0a constant-current/constant-voltage battery charger step-down charger that allows charging during computer operation and with input current limiting prevents wall-adapter overload lt1513 sepic constant-current/constant-voltage battery charger step-up/step-down charger for up to 2a current related parts

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